As a technique for measuring a phase fluctuation (jitters) of a pulse signal such as a data signal or a clock signal, there is known a time interval measuring method for measuring an interval time between a rising portion and a falling portion of the pulse signal.
In such a time interval measuring method, it is demanded that the measurement thereof should be more precise along with an increase in a data transmission rate in recent years.
For example, since an interval between bits of a digital signal of 1 Gbps is 1 nsec, at least 100 psec of the measurement precision is required in the case of the time interval measurement method in which this digital signal is set as an object of measurement.
As the most simple method for measuring an input interval time of a pulse signal, there is provided a so-called period measurement method for counting a high frequency signal having a stable frequency from an input timing (a timing of a rise and a fall) of a pulse signal to be measured up to the next input timing thereof.
However, in order to measure the period in a precision of 100 psec as described above, it is required to prepare a counter which directly counts a high frequency signal having a frequency of tens of GHz or more without dividing the frequency. Under these circumstances, it is difficult to adopt such a measurement method.
As a method for solving this problem, there is known a technique for measuring in a high precision an input interval time of a pulse signal with a combination of the period measurement method and a phase measurement method for measuring a phase variation of an analog signal having a predetermined period.
FIG. 12 is a view showing a configuration of a conventional time interval measuring apparatus 10 in which the period measurement method and the phase measurement method are combined.
A reference signal generator 11 in the time interval measurement apparatus 10 generates an analog reference signal r(t) having a predetermined frequency number fr (a period Tr) and a clock signal c which is synchronized in phase with the reference signal r(t) at the same frequency.
As the reference signal r(t), a trapezoid wave is used. In the trapezoid wave, for example, as shown in (a) of FIG. 13, when one period of the reference signal r(t) is set to Tr, the voltage is monotonously increased from 0 to V with a constant inclination within the range of the first (Tr/4) period (a phase of 0 to π/2); the voltage is set to a definite level of V within the range of the next (Tr/4) period (a phase of π/2 to π); the voltage is monotoneously decreased from V to 0 with a constant inclination within the range of the next (Tr/4) period (a phase of π to 3π/2); and furthermore, the voltage is set to a definite level of 0 within the range of the next (Tr/4) period (phase of 3π/2 to 2π).
A phase shifter 12 to which this reference signal r(t) is input generates a first analog signal i(t) and a second analog signal q(t) which have the same waveform as that the reference signal r(t) and which have phases mutually different through 90 degrees as shown in (b) and (c) of FIG. 13. The phase shifter 12 outputs the first analog signal i(t) to a first analog/digital (A/D) converter 13, and outputs the second analog signal q(t) to a second A/D converter 14.
Here, as shown in (a) and (b) of FIG. 13, the first analog signal i(t) and the reference signal r(t) are made to have the same phase.
Furthermore, it has been explained that each of the A/D converters 13 and 14 incorporates a track hold (a sample hold) circuit (not shown). In the case where an A/D converter which incorporates no track hold circuit, an independent track hold circuit is provided on the front stage thereof.
To both A/D converters 13 and 14, a pulse signal p to be measured is commonly input.
The first A/D converter 13 performs sampling of the first analog signal i(t) at an input timing tn (for example, a rise timing) of a pulse signal p shown in (d) of FIG. 13, and converts its analog sample value i(tn) into a digital sample value I(tn) to output the value as shown in (e) of FIG. 13 (n=1, 2, 3, . . . ).
Further, the second A/D converter 14 performs sampling of the second analog signal q(t) at the input timing tn (for example, the rise timing) of the pulse signal p shown in (d) of FIG. 13 in the same manner, and converts its analog digital sample value Q(tn) into a digital sample value Q(tn) to output the value as shown in (f) of FIG. 13 (n=1, 2, 3, . . . ).
The instantaneous phase calculator 15 determines an instantaneous phase φ(tn) in one period of the reference signal r(t) at the time of the input of the pulse signal (p) based on the digital sample values I(tn) and Q(tn) which are output from both the A/D converters 13 and 14, and outputs the instantaneous phase as shown in (g) of FIG. 13.
Here, the first analog signal i(t) and the second analog signal q(t) have the same waveform as the reference signal r(t), and have a trapezoid-shaped waveform in which the inclination thereof changes in each of π/2 as described above. Besides, since the two analog signals have phases which are shifted for each of π/2, the phase can be specified from the two digital sample values I(tn) and Q(tn).
That is, like the case of time t1 of FIG. 13D, in the case where the digital sample value I(tn) is set within the range of 0 or more and less than V, and the digital sample value Q(tn) is set to 0, the instantaneous phase φ(tn) is set to the range of 0 or more and less than π/2, and the instantaneous phase can be determined from the following calculation:φ(tn)=(π/2)·I(tn)/V.
In addition, when the digital sample value I(tn) is set to V and the digital sample value Q(tn) is set to the range of 0 or more and less than V, the instantaneous phase φ(tn) is set to the range of π/2 or more and less than π, and can be determined from the following calculation:φ(tn)=(π/2)+(π/2)·Q(tn)/V.
Further, like the case of time t2 as shown in (d) of FIG. 13, when the digital sample value I(tn) is set to the range of V or less and more than 0 and the digital sample value Q(tn) is set to V, the instantaneous phase φ(tn) is set to the range of π or more and less than 3π/2, and the instantaneous phase can be determined from the following calculation:φ(tn)=π+(π/2)·[V−I(tn)]/V.
Furthermore, when the digital sample value I(tn) is set to 0 and the digital sample value Q(tn) is set to the range of V or less and more than 0, the instantaneous phase φ(tn) is set to the range of 3π/2 or more and less than 2π, and the instantaneous phase can be determined from the following calculation:φ(tn)=(3π/2)+(π/2)·[V−Q(tn)]/V.
On the other hand, the counter 17 counts (adds and counts) a clock signal c which is output as shown in (h) of FIG. 13 from the reference signal generator 11, and subsequently outputs its count value M as shown in (i) of FIG. 13.
The interval time calculation unit 18 receives the count value M output from the counter 17, the instantaneous phase φ(tn) output from the instantaneous phase calculation unit 15, and the pulse signal p, and calculates an input interval time T of the pulse signal p.
For example, as shown in (d) of FIG. 13, in the case where an interval T of input timings t1 and t2 of the pulse signal p is determined, a difference Δφ (a radian) between instantaneous phases φ(t1) and φ(t2) obtained at the respective timings is determined from the following calculation:Δφ=φ(t2)−φ(t1).
In addition, a difference ΔM (a stepping number of the counter 17) between a count value M(t1) at t1 and a count value M(t2) at t2 is determined from the following calculation:ΔM=M(t2)−M(t1).(In the example shown in (i) of FIG. 13, the following mathematical expression is established: ΔM=u+2−u=2).
Then, the interval T is calculated from the following calculation:T=(1/fr)·(ΔM+Δφ/2π).
A precision of the interval T calculated by the time interval measuring apparatus 10 having the configuration is determined with a precision of the phase difference Δφ while a precision of the phase difference Δφ depends on the bit numbers of the A/D converters 13 and 14.
For example, in the case where each of the A/D converters 13 and 14 is of a 14 bits type, voltages of the analog signals i(t) and q(t) ranging from 0 to the maximum voltage V are represented in values from all bits 0 to all bits 1, and a reference signal frequency fr is 10 MHz, a time required for the voltage of the trapezoid waveform to change from 0 to V becomes 25 nsec.
Consequently, the time resolution during the time becomes about 25/16000 (nsec)=25/16 (psec). In theory, the precision of several psec can be realized.
Incidentally, as described above, the following patent document 1 describes a technique in which a reference signal r(t) having an analog trapezoid-shaped wave is divided into the analog signals i(t) and q(t) having different phases through 90 degrees by a phase shifter 12, the analog signals i(t) and q(t) are input to the A/D converters 13 and 14, respectively to perform sampling of the analog signals i(t) and q(t) with a pulse signal p to be measured, thereby determining an input interval time of the pulse signal p to be measured based on the digital sample values I(tn) and Q(tn).
Patent Document 1: Jpn. Pat. Appln. KOKAI Publication No. 5-215873
However, the measurement precision of the several psec is a theoretical value. With an actual time interval measuring apparatus 10, it is extremely difficult to measure the input interval time of the pulse signal p to be measured accurately and stably with the precision depending on a diagonal error of the phase shifter 12, a characteristic difference and environmental change of the A/D converters 13 and 14.
That is, in the case of the 90 degrees hybrid type which is generally used as an analog phase shifter 12, a diagonal error (a phase error and an amplitude error) is present. Besides, the error largely changes with environmental changes such as the frequency and temperatures.
Therefore, even if a circuit constant in the phase shifter 12 is adjusted at a certain frequency and under a certain environment to obtain a desired precision, it is impossible to maintain the precision over a long time.
Further, since the reference signal r(t) having the trapezoid-shaped wave includes many high frequency components in addition to its fundamental wave components, it is demanded that the phase shifter 12 which performs a phase shift process without deforming such a reference signal r(t) having the trapezoid-shaped wave has a wideband characteristic which is several times wider than the frequency fr of the fundamental wave.
However, in actuality, it is substantially impossible to prevent a change of the error characteristic by the phase shifter 12 having such a wideband characteristic.
When an example of the numeric value is shown, a phase error of one degree in the phase shifter 12 at the frequency fr of the reference signal corresponds to a measurement error of 1/(fr·360) sec p-p, and a measurement error when the frequency fr is 10 MHz becomes 277 psec p-p.
On the other hand, in order to set the measurement error to 100 psec or less, it is required to set the phase error of the phase shifter 12 to 0.3 degree or less in a wide frequency band, which is impossible to realize.
Furthermore, the A/D converters 13 and 14 respectively have individual delay characteristics, gain characteristics, and direct current offset characteristics. Depending on differences in these characteristics, a phase error, an amplitude error and a direct current offset error may occur between the digital sample values I and Q.
Generally, a delay time of a sampling process of an A/D converter is several hundreds of psec on the shortest level. It is extremely difficult to allow a difference in delay time between two A/D converters 13 and 14 to a level on the order of 100 psec or less.
In addition, even if the difference in delay time is eliminated by a delay process with respect to a pulse signal p which is input to the side of one of the A/D converters, it is impossible to obtain a long time precision owing to the characteristic change resulting from the temperature of the delay process unit.